Power amplifier open loop current clamp

ABSTRACT

Various implementations include circuits, devices and/or methods that provide open loop current limiting power amplifiers and the like. In some implementations, an open loop current clamp includes a trim module to provide a control value and a limiting source having respective input and output terminals. The input terminal is coupled to the trim module to receive the control value. The output terminal coupled to a control terminal of the first transistor to provide a limiting electrical level produced in response to the control value by the limiting source. The limiting electrical level substantially setting a first mode of operation for the first transistor such that the current draw of the first transistor is substantially determined by the first mode of operation and the limiting electrical level such that a voltage at an output terminal of the first transistor exerts reduced influence on the current draw.

RELATED APPLICATIONS

This application claims the benefit of U.S. Provisional PatentApplication No. 61/860,548, filed on Jul. 31, 2013, and which isincorporated by reference herein.

TECHNICAL FIELD

The present disclosure relates to electronic circuits, and inparticular, to systems, methods and devices configured to controlcurrent draw by a power amplifier.

BACKGROUND

Power amplifiers are widely used in various communication networks toset the transmission power level of an information-bearing signaltransmitted by one device to another device. For example, poweramplifiers are used to set the pulse energy emitted by pulsed lasers inoptical communication networks. Power amplifiers are also used in theradio frequency (RF) front end components of wireless carrier networkdevices—such as base stations, repeaters, and mobile client devices(e.g. mobile phones, smartphones, tablet computers, etc.)—to set thepower level of a wireless signal transmitted through an antenna. Poweramplifiers are also used in local area networks of homes and offices tosupport both wired and wireless connectivity of servers, computers,laptops, and peripheral devices such as photocopiers and printers.

In a mobile device relying on a battery, managing the operation of apower amplifier is of interest because the efficiency of the poweramplifier is often a significant factor in the overall efficiency of theRF front end, and in turn, the battery life of the mobile device.Additionally, in part because of the high power levels at which a poweramplifier operates relative to the other components in the RF front end,a power amplifier can cause device failure when the power amplifier isnot prevented from drawing excessive current and/or causing a voltagespike, both of which can reach destructive levels.

In order to set the operating conditions for efficient operation, apreferred current draw of a power amplifier can be, in part, initiallyset by impedance matching the power amplifier to the antenna (or othertransmission load), so that the power amplifier operates within adesigned quiescent range set by bias circuitry. However, even when apower amplifier is impedance matched to an antenna, the power amplifiermay draw excessive current and/or cause a voltage spike when subjectedto unanticipated antenna load conditions. Consequently, battery life isreduced and device failure may occur if the excessive current and/orcause a voltage spike reach destructive levels.

BRIEF DESCRIPTION OF THE DRAWINGS

So that the present disclosure can be understood by those of ordinaryskill in the art, a more detailed description may be had by reference toaspects of some illustrative implementations, some of which are shown inthe accompanying drawings.

FIG. 1 is a schematic diagram of a power amplifier operating controlconfiguration according to some implementations.

FIG. 2 is a schematic diagram of a closed loop current clamp circuitprovided to control the operation of a power amplifier according to someimplementations.

FIG. 3 is a schematic diagram of a closed loop current clamp circuitprovided to control the operation of a power amplifier according to someimplementations.

FIG. 4 is a schematic diagram of an open loop current clamp circuitprovided to control the operation of a power amplifier according to someimplementations.

FIG. 5 is a schematic diagram of a current trim circuit included in theopen loop current clamp circuit of FIG. 4.

FIG. 6 is a flowchart of an implementation of a method of clamping thecurrent drawn by a power amplifier according to some implementations.

FIG. 7 is a performance diagram showing a power amplifier final stagebeta versus load angle under current limits imposed by an implementationof the open loop current clamp circuit of FIG. 4.

FIG. 8 is a performance diagram showing a power amplifier final stagebeta versus temperature under current limits imposed by animplementation of the open loop current clamp circuit of FIG. 4.

FIG. 9 is a performance diagram showing a power amplifier final stagecollector current versus load angle under current limits imposed by animplementation of the open loop current clamp circuit of FIG. 4.

FIGS. 10A-10C are schematic diagrams of different integrated circuitimplementations of the open loop current clamp circuit of FIG. 4.

FIG. 11 is a schematic diagram of an implementation of a moduleincluding the open loop current clamp circuit of FIG. 4.

FIG. 12 is a schematic diagram of an implementation of a wireless deviceincluding the open loop current clamp circuit of FIG. 4.

In accordance with common practice various features shown in thedrawings may not be drawn to scale, as the dimensions of variousfeatures may be arbitrarily expanded or reduced for clarity. Moreover,the drawings may not depict all of the aspects and/or variants of agiven system, method or apparatus admitted by the specification.Finally, like reference numerals are used to denote like featuresthroughout the drawings.

SUMMARY

Various implementations of circuits, methods and devices within thescope of the appended claims each have several aspects, no single one ofwhich is solely responsible for the attributes described herein. Withoutlimiting the scope of the appended claims, some prominent features aredescribed. After considering this disclosure, and particularly afterconsidering the section entitled “Detailed Description,” one willunderstand how the aspects of various implementations enable open loopcurrent limiting for power amplifiers.

Some implementations include an open loop current clamp to restrict thecurrent draw of a first transistor. In some implementations, the openloop current clamp includes a trim module to provide a control value anda limiting source having respective input and output terminals. Theinput terminal is coupled to the trim module to receive the controlvalue. The output terminal coupled to a control terminal of the firsttransistor to provide a limiting electrical level produced in responseto the control value by the limiting source. The limiting electricallevel substantially setting a first mode of operation for the firsttransistor such that the current draw of the first transistor issubstantially determined by the first mode of operation and the limitingelectrical level such that a voltage at an output terminal of the firsttransistor exerts reduced influence on the current draw.

Some implementations include a power amplifier having an open loopcurrent clamped first transistor. In some implementations, the open loopcurrent clamp includes a trim module to provide a control value and alimiting source having respective input and output terminals. The inputterminal is coupled to the trim module to receive the control value. Theoutput terminal coupled to a control terminal of the first transistor toprovide a limiting electrical level produced in response to the controlvalue by the limiting source. The limiting electrical levelsubstantially setting a first mode of operation for the first transistorsuch that the current draw of the first transistor is substantiallydetermined by the first mode of operation and the limiting electricallevel such that a voltage at an output terminal of the first transistorexerts reduced influence on the current draw.

Some implementations include a power amplifier module. The poweramplifier module, in some implementations, includes a packagingsubstrate configured to receive a plurality of components; a poweramplifier circuit provided in a die included on the packaging substrate,the power amplifier including a first transistor having a controlterminal and an output terminal; a trim module to provide a controlvalue; a limiting source having respective input and output terminals,the input terminal coupled to the trim module to receive the controlvalue, and the output terminal coupled to the control terminal of thefirst transistor to provide a limiting electrical level produced inresponse to the control value by the limiting source, the limitingelectrical level substantially setting a first mode of operation for thefirst transistor such that the current draw of the first transistor issubstantially determined by the first mode of operation and the limitingelectrical level such that a voltage at the output terminal of the firsttransistor exerts reduced influence on the current draw.

Some implementations include a radio frequency (RF) device. In someimplementations, the RF device includes a transceiver configured toprocess RF signals; an antenna in communication with the transceiverconfigured to facilitate transmission of an amplified RF signal; and apower amplifier module connected to the transceiver and configured togenerate the amplified RF signal. In some implementations, the poweramplifier module includes a first transistor, a trim module to provide acontrol value, and a limiting source having respective input and outputterminals, the input terminal coupled to the trim module to receive thecontrol value, and the output terminal coupled to a control terminal ofthe first transistor to provide a limiting electrical level produced inresponse to the control value by the limiting source, the limitingelectrical level substantially setting a first mode of operation for thefirst transistor such that the current draw of the first transistor issubstantially determined by the first mode of operation and the limitingelectrical level such that a voltage at an output terminal of the firsttransistor exerts reduced influence on the current draw.

Some implementations include a method of clamping the current of a firsttransistor. In some implementations, the method includes producing acontrol value using a trim module to compensate for at least one oftemperature and manufacturing process variations; generating a limitingelectrical level using the control value; and, applying the limitingelectrical level to a control terminal of a first transistor.

DESCRIPTION

Numerous details are described herein in order to provide a thoroughunderstanding of the example implementations illustrated in theaccompanying drawings. However, the invention may be practiced withoutmany of the specific details. Well-known methods, components, andcircuits have not been described in exhaustive detail so as not tounnecessarily obscure more pertinent aspects of the implementationsdescribed herein.

As noted above, in order to set the operating conditions for efficientoperation, a preferred current draw of a power amplifier can be, inpart, initially set by impedance matching the power amplifier to theantenna (or other transmission load), so that the power amplifieroperates within a designed quiescent range set by bias circuitry. Thoseskilled in the art will appreciate that impedance matching refers tomatching the output impedance of a circuit that transmits a signal andthe input impedance of a circuit that receives the signal in order toobtain efficient power transfer. When a power amplifier and atransmission load (e.g. an antenna) are impedance matched, efficientpower transfer may be obtained, and the generation of a reflected wavefrom the transmission load may be suppressed. Impedance matching alsosets the preferred voltage standing wave ratio (VSWR) between poweramplifier and the transmission load. Those skilled in the art willappreciate from the present disclosure that VSWR may be brieflydescribed as a ratio of a maximum value of a voltage standing wave to aminimum value of the voltage standing wave. Moreover, those skilled inthe art will appreciate that a voltage standing wave on a transmissionpath between the power amplifier and the transmission load is thecombination of a transmission signal emitted by the power amplifier, anda reflected signal from the transmission load to the power amplifier.

Impedance matching is often fixed such that the impedances match over aparticular range of frequencies, load angles, temperatures,environmental conditions and physical attributes of the impedancematched components. However, during operation one or more of the variousfactors may change—such as changes in the operating environment(temperature, humidity, etc.), deterioration due to aging, and/orphysical damage—that in turn changes the effective input impedance ofthe transmission load temporarily or permanently. Changes to theeffective impedance of the transmission load cause an impedance mismatchbetween the transmission load and power amplifier. In turn, a reflectedvoltage wave is generated, and the VSWR between power amplifier and thetransmission load shifts from the designed range. When the VSWR movesoutside of the designed range the power amplifier may operate outside ofthe designed quiescent range and draw excessive current and/or cause avoltage spike. Depending on the magnitude of the VSWR change, the poweramplifier may draw an excessive current and/or cause a voltage spikethat is destructive to the power amplifier or components situated nearthe power amplifier.

In other words, even when a power amplifier is impedance matched to anantenna, the power amplifier may draw excessive current and/or cause avoltage spike when subjected to unanticipated antenna load conditions.Consequently, battery life is reduced and device failure may occur ifthe excessive current and/or cause a voltage spike reach destructivelevels.

FIG. 1, for example is a schematic diagram of an example power amplifier(PA) operating configuration 100 according to some implementations.While pertinent features are illustrated, those skilled in the art willappreciate from the present disclosure that various other features havenot been illustrated for the sake of brevity and so as not to obscuremore pertinent aspects of the example implementations disclosed herein.To that end, as a non-limiting example, the PA operating configuration100 includes a PA 102, a PA control module 104, a matching circuit 108and an antenna 110. The PA control module 104 is coupled to the PA 102through control line 106 in order to provide at least one control signalto the PA 102. The PA 102 is coupled to receive a RF input signal(RF_(in)) through coupling capacitor 103 (C), and provide an amplifiedRF output signal (RF_(out)) to the antenna 110 through the matchingcircuit 108. The PA 102 draws a DC operating current from the voltagesupply 105.

In some implementations, the capacitor 103 is implemented as, forexample, a single capacitor to block undesired DC currents and/orvoltages from the input node of the PA 102, while allowing the transferof RF signals. In some implementations, the capacitor 103 is implementedas a multi-component capacitive network. Alternatively, although FIG. 1shows the capacitor 103, it will be understood that one or more featuresof the present disclosure can also be implemented in a system or devicewithout such a similarly situated capacitance. Similarly, in someimplementations, a RF choke inductor (not shown) can be placed in seriesbetween the voltage supply 105 and the PA 102 to prevent RF energytransfer between the voltage supply 105 and the PA 102.

In order to maintain the operating conditions for an efficient range ofoperation, a preferred current draw of the PA 102 can be in partinitially maintained by impedance matching the PA 102 to the antenna110. In other words, the matching circuit 108 impedance matches the PA102 to the antenna 110 so that the PA 102 operates within the designedcurrent range. Those skilled in the art will appreciate from the presentdisclosure that a matching circuit generally includes a combination ofcapacitive and inductive components, and/or transmission lines, etc. Insome implementations, the PA 102 is not conjugately matched to the load,as is done with a small signal amplifier. For a large signal PA, theoutput match presents a preferred load line to the PA collector in orderto improve voltage and current excursions, and thus, the resulting powerdelivered to the load. When the PA 102 and the antenna 110 are impedancematched the generation of a reflected wave from the antenna 110 issuppressed, which encourages efficiency.

The impedance matching circuit 108 also sets the preferred voltagestanding wave ratio (VSWR) between the PA 102 and the antenna 110. Insome implementations, the preferred standing wave is reduced tonegligible, if not non-existent, levels. In some implementations, thepreferred standing wave occurs when the min-to-max voltage ratio isclose to one within a level of tolerance. Those skilled in the art willappreciate from the present disclosure that VSWR is generally describedas a ratio of a maximum value of a voltage standing wave to a minimumvalue of the voltage standing wave. Moreover, those skilled in the artwill appreciate that a voltage standing wave on a transmission pathbetween the PA 102 and the antenna 110 is the combination of atransmission signal emitted by the PA 102 and a reflected signal fromthe antenna 110.

Impedance matching is often fixed such that the matching circuit 108matches impedances over a particular range of frequencies, load angles,temperatures, environmental conditions and various physical attributesof the impedance matched components. However, during operation one ormore of the aforementioned factors may change, which in turn causes animpedance mismatch between the components temporarily or permanently.Such changes include, for example, changes in the operating environment(temperature, humidity, etc.), deterioration due to aging, and/orphysical damage to the antenna 110, which changes the effective inputimpedance, Z_(in), of the antenna 110. Changes to the effective inputimpedance, Z_(in), of the antenna 110 cause an impedance mismatchbetween the antenna 110 and the PA 102. In turn, a reflected voltagewave is generated, and the VSWR between the PA 102 and the antenna 110shifts from the designed range. When the VSWR moves outside of thedesigned range the PA 102 may operate outside of the designed quiescentrange and draw excessive current and/or cause a voltage spike. Dependingon the magnitude of the VSWR change, the excessive current and/orvoltage spike may be destructive to the PA 102 and/or RF front endcomponents situated near the PA 102.

In other words, even when a power amplifier is initially impedancematched to an antenna, the power amplifier may draw excessive currentand/or cause a voltage spike when subjected to unanticipated antennaload conditions that affect the impedance match. Consequently, batterylife is reduced and device failure may occur if the excessive currentand/or voltage spike reach destructive levels. Accordingly, a controlcircuit, such as the PA control circuit 104 shown in FIG. 1, is used tolimit the current draw by the power amplifier (e.g. the PA 102).

The various implementations described herein include systems, methodsand/or circuit-based devices provided to limit current draw by a poweramplifier from a power supply. Numerous details are described herein inorder to provide a thorough understanding of the example implementationsillustrated in the accompanying drawings. However, the invention may bepracticed without many of the specific details. Well-known methods,components, and circuits have not been described in exhaustive detail soas not to unnecessarily obscure more pertinent aspects of theimplementations described herein.

To that end, FIG. 2 is a schematic diagram of a closed loop currentclamp circuit 200 provided to control the operation of a power amplifieraccording to some implementations. The closed loop current clamp circuit200 illustrated in FIG. 2 is similar to and adapted from the PAoperating configuration 100 illustrated in FIG. 1. Elements common toeach include common reference numbers, and only the differences betweenFIGS. 1 and 2 are described herein for the sake of brevity. Moreover,while certain specific features are illustrated, those skilled in theart will appreciate from the present disclosure that various otherfeatures have not been illustrated for the sake of brevity and so as notto obscure more pertinent aspects of the example implementationsdisclosed herein.

Specifically, the closed loop current clamp circuit 200 includes acurrent clamp 204 arranged in a closed feedback loop with the PA 100.The current clamp 204 includes a current sense circuit 204 a and acompensation circuit 204 b. The current sense circuit 204 a is coupledto sense the current drawn by the PA 102 from the voltage supply 105. Insome implementations using bipolar junction transistors (BJTs), thecurrent draw by the PA 102 is often referred to as at least one of thecollector and emitter currents of a respective BJT included in the PA102. The current sense circuit 204 a is also coupled to the compensationcircuit 204 b to provide an indication of the sensed current. Forexample, in some implementations, the indication of the sensed currentis an electrical signal proportional to the sensed current, such as avoltage or current.

The compensation circuit 204 b provides the control signal on controlline 106 to the PA 102. The control signal causes the PA 102 to adjustthe current draw from the voltage supply 105. In some implementations,the control signal is at least one of a new bias current and voltage.Additionally and/or alternatively, in some implementations, the controlsignal is a representation of a change to at least one of a bias currentand voltage to be made by power control circuitry (see FIG. 12)associated with the PA 102.

FIG. 3 is a schematic diagram of a closed loop current clamp circuit 300provided to control the operation of a power amplifier according to someimplementations. The closed loop current clamp circuit 300 illustratedin FIG. 3 is similar to and adapted from the closed loop current clampcircuit 200 illustrated in FIG. 2. Elements common to each includecommon reference numbers, and only the differences between FIGS. 2 and 3are described herein for the sake of brevity. Moreover, while certainspecific features are illustrated, those skilled in the art willappreciate from the present disclosure that various other features havenot been illustrated for the sake of brevity and so as not to obscuremore pertinent aspects of the example implementations disclosed herein.

Specifically, the closed loop current clamp circuit 300 includes a bondwire 311 in series with a bias choke inductor 314 (L_(CC)) between thevoltage supply 105 and a final stage of the power amplifier 302. In someimplementations, the bond wire 311 includes two or more parallel bondwires. As an approximation, the bond wire 311 is modeled as havingparasitic inductive and resistive components, shown in FIG. 3 asinductor 311 a (L_(BW)) and parasitic resistor 311 b (R_(PAR)).Additionally and/or alternatively, in some implementations, one or moresurface mount resistors (not shown) are coupled in series between thechoke inductor 314 and the voltage supply 105.

The closed loop current clamp circuit 300 also includes a current clamp304. The current clamp 304 includes a reference resistor 310 (R_(REF))and a reference current source 307 (I_(REF)) connected in series betweenthe voltage supply 105 and ground. The current clamp 304 also includesan op-amp 304 a arranged to compare the voltage drop, V_(REF), acrossthe reference resistor 310 to the voltage drop, V_(SNS), across the bondwire 311. To that end, one input terminal of the op-amp 304 a is coupledto node 308 between the reference resistor 308 and reference currentsource 307, and another input terminal of the op-amp 304 a is coupled tonode 306 between the bond wire 311 and choke inductor 314. An outputterminal of the op-amp 304 a is coupled to a bias compensation circuit304 b, which is, in turn, coupled to the PA final stage 302 via controlline 106 to deliver a new bias voltage and/or current. The op-amp 304 aand bias compensation circuit 304 b thus form a closed loop integratorwith the PA final stage 302.

In operation, an indication of the current, I_(PA), drawn by the PAfinal stage 302 is a voltage error signal proportional to the differencebetween the current I_(PA) and the reference current I_(REF). Morespecifically, a DC voltage V_(SNS) is developed across the bond wire(s)311 (and/or resistors) that is proportional to the current drawn I_(PA)by the PA final stage 302. In other words, the DC voltage V_(SNS) servesas a proxy for adjusting current draw by the PA final stage 302.

The voltage V_(SNS) is continuously compared to the reference voltageV_(REF) across the reference resistor 310 using the op-amp 304 a whichgenerates and provides a voltage error to the bias compensation circuit304 b. Under high current conditions, the bond wire voltage V_(SNS)exceeds the reference voltage V_(REF), and the PA driver collectorvoltage is reduced, decreasing the RF drive to the PA final stage 302and decreasing the final stage collector current I_(PA). This processcontinues until the closed loop integrator voltage error generated bythe op-amp 304 a is driven to zero at which point the final collectorcurrent is approximately equal to the desired current limit thresholddetermined by the integrator reference voltage V_(REF).

In other words, the current draw by the PA final stage 302 is initiallypermitted to deviate away from the preferred designed range before theclosed loop integrator reacts to reduce the current draw back to withinan acceptable range. The time it takes to reduce the current back towithin an acceptable range may lead to performance risks in someimplementations. For example, because the current draw by the PA finalstage is permitted to deviate substantially, in some instances, thecurrent I_(PA) may reach potentially destructive levels before thecurrent is reduced by the closed loop integrator. Additionally, withthis approach, in some implementations precise routing of the two sensePCB traces and additional RF decoupling components are used to avoidbias controller RF rectification which may result in clamp thresholdvariation with power and frequency.

By contrast, FIG. 4 is a schematic diagram of an open loop current clampcircuit 400 provided to control the operation of a power amplifieraccording to some implementations. Briefly, the open loop current clamp400 includes a pre-clamped current source that substantially reduces theability of a power amplifier to draw excessive current from the voltagesupply and substantially removes the need for a reactive mechanism tocontrol current draw. In some implementations, the open loop currentclamp includes a pre-clamped current source, and a current trim module.The current output of the pre-clamped current source is limited to aprogrammable current level set by the current trim module. To that end,the open loop current clamp circuit 400 of FIG. 4 includes a voltagesupply 405, a reference current source 403 (I_(REF)), a limiting currentsource 404 (I_(b) _(_) _(clamp), i.e., the pre-clamped source), a trimmodule 420, and a PA final stage 402 an example active load.

The PA final stage 402 includes a first transistor 416 (Q₁). In someimplementations, first transistor 416 (Q₁) is a single transistor.Additionally and/or alternatively, in some implementations, the firsttransistor 416 (Q₁) comprises an array of transistors that can beschematically represented by Q₁. In various implementations, Q₁ is oneof a bipolar junction transistor (BJT), and a heterojunction bipolartransistor (HBT). For the sake of example only, and without loss ofgenerality, the first transistor 416 (Q₁) will be described as a BJThereinafter. An RF choke inductor 415 is placed in series between thevoltage supply 405 and the first transistor 416 (Q₁) to prevent RFenergy transfer between the voltage supply 405 and the first transistor416 (Q₁). A DC block capacitor 412 (C₂) is coupled to the base of thefirst transistor 416 (Q₁) to block undesired DC currents and/or voltagesfrom the base of the first transistor 416 (Q₁), while allowing thetransfer in of an RF signal from a PA drive stage (not shown).

The quiescent DC biasing of the first transistor 416 (Q₁) is provided bya current mirror, and the combination of the limiting current source 404and current trim module 420. The current mirror includes a secondtransistor 411 (Q₂), a beta helper 410 (M₁) and the aforementionedreference current source 403. The collector (i.e., an input terminal) ofthe second transistor 411 (Q₂) is coupled to the output of the referencecurrent source 403. The base of the second transistor 411 (Q₂) iscoupled to the base of the first transistor 416 (Q₁). The base of a BJTcan be used control the operation of the BJT. Accordingly, as usedherein, the base is referred to as a control terminal. Thus, withcontinued reference to FIG. 4, in other words, the control terminal ofthe second transistor 411 (Q₂) is coupled to the control terminal of thefirst transistor 416 (Q₁). The current mirror includes a resistor 413(R₁) and a capacitor 414 (C₁) in series between the input terminal andcontrol terminal of the second transistor 411 (Q₂) to provide stability.

In some implementations, the first transistor 416 (Q₁) and the secondtransistor 411 (Q₂) are together characterized by a characterized by adevice area ratio. The device area ratio is used to determine in partthe current draw by the first transistor 416 (Q₁) relative to the outputof the reference current source 403.

The beta helper 410 (M₁) includes a MESFET transistor having first,second and third terminals. Although a MESFET transistor is shown, insome implementations the beta helper 410 (M₁) is a MOS or JFETtransistor. The first terminal (i.e., the gate) of the beta helper 410(M₁) is coupled to the output of the reference current source 403. Thesecond terminal (i.e., the drain) of the beta helper 410 (M₁) is coupledto the output of the limiting current source 404. The third terminal(i.e., the source) of the beta helper 410 (M₁) is coupled to the controlterminal (i.e, the base) of the first transistor 416 (Q₁).

In operation, the quiescent collector current of the first transistor416 (Q₁) is can be approximated according to equation (1):I _(CQ1) =A*I _(REF)  (1)where A is the device area ratio (Q₂/Q₁) between the first and secondtransistors Q₁ and Q₂.

The collector current, I_(CQ1), and the base current, I_(BQ1), can beapproximated by equation (2):I _(CQ1) I=β(temperature,Process)*I _(BQ1)  (2)where β is the current gain of the first transistor 416 (Q₁), whichvaries as a function of temperature and manufacturing processcharacteristics.

In some implementations, the collector current, I_(CQ1), of the firsttransistor 416 (Q₁) is clamped by fixing the base current, I_(BQ1), ofthe first transistor 416 (Q₁) using the combination of the limitingcurrent source 404 and current trim module 420. More specifically, thecurrent trim module 420 provides a control value to the limiting currentsource 404. The limiting current source in turn creates a limitingelectrical level, current I_(b) _(_) _(clamp), in response to receivingthe control value. The current I_(b) _(_) _(clamp), when provided to thebase of the first transistor 416 (Q₁), ensures that the first transistor416 (Q₁) remains in an active-mode of operation, where for BJTs, thecollector current is substantially determined by the base current, andsubstantially independent of the voltage at the collector. That is, thecurrent draw by the first transistor 416 (Q₁) is less susceptible tovoltage changes at an output terminal of the first transistor 416 (Q₁).More generally, limiting electrical level substantially sets a firstmode of operation (e.g., active mode for BJTs) for the first transistorsuch that the current draw of the first transistor is substantiallydetermined by the first mode of operation and the limiting electricallevel such that a voltage at an output terminal of the first transistorexerts reduced influence on the current draw.

By contrast, without the combination of the limiting current source 404and current trim module 420, a standard current mirror (not shown) ismore susceptible to voltage variations at the collector (or drain) ofthe output transistor. As such, the load (e.g. an antenna) coupled tothe output transistor is typically required to ensure that the outputtransistor operates in the active-mode for a BJT. As, noted above,impedance matching can be used to match the output of impedance of theoutput transistor to the input impedance of the load in order to set theinitial operating conditions. However, as also noted above, impedancematching is often fixed, and thus, cannot alone ensure the desiredoperating conditions over the life of the device. An appreciable changein the VSWR at the output terminal of the output transistor can changethe current draw of the output transistor, and in turn, the otherportions of the current mirror.

With continued reference to FIG. 4, to compensate for temperature andpart-to-part process variation and to meet an overall +/−5% currentclamp threshold variation specification, the reference voltage istrimmed at production test using programmable resistive fuses. FIG. 5 isa schematic diagram of the combination of the limiting current source404 and the current trim module 420 included in FIG. 4.

The limiting current source 404 includes an adjustable current mirror.The adjustable current mirror includes a first PMOS transistor 512 (M₃).The source of the first PMOS transistor 512 (M₃) is coupled to thevoltage supply 405, and the drain is coupled to node 544. The adjustablecurrent mirror also includes a plurality of transistors 511 a, 511 b,511 c, 511 d. The respective sources of the plurality of transistors 511a, 511 b, 511 c, 511 d are each coupled to the voltage supply 405. Therespective drains are each coupled to one another, and ultimately to thebase of the first transistor 416 (Q₁) of FIG. 4 to supply the limitingcurrent level, I_(BQ1) or I_(b) _(—clamp) . The gate of the transistor511 a is coupled to the gate of the first PMOS transistor 512 (M₃) inorder to provide the basis of the adjustable current mirror. The gate ofthe transistor 511 b is selectively connectable to the gate of the firstPMOS transistor 512 (M₃) using switch 513 a in order to provide atwo-fold (2×) increase in the reference current of the current mirror.Similarly, the gate of the transistor 511 c is selectively connectableto the gate of the first PMOS transistor 512 (M₃) using switch 513 b inorder to provide a four-fold (4×) increase in the reference current ofthe current mirror. Similarly, the gate of the transistor 511 d isselectively connectable to the gate of the first PMOS transistor 512(M₃) using switch 513 c in order to provide an eight-fold (8×) increasein the reference current of the current mirror. In operation, theadjustable current mirror allows for dynamic adjustment of the limitingcurrent after trim fuses resistors have been set through the operationof the switches 513 a, 513 b, 513 c.

In some implementations, the limiting current source 404 includes anon-adjustable current source including a pair of gate-coupled PMOStransistors, with the respective source terminal of each coupled to thevoltage supply 405, and the drain of one is coupled to the base of thefirst transistor 416 (Q₁) of FIG. 4 to supply the limiting currentlevel, I_(BQ1) or I_(b) _(—clamp) .

The current trim module 420 includes a voltage reference source 520, anadjustable load 540, and an op-amp 530 serving as a control valuegenerator. The adjustable load 540 is coupled to sink current from thedrain of M₃. The output terminal of the op-amp 530 is coupled to thegates of M₂ and M₃ to set the limiting current level, I_(BQ1). In otherwords, the output of the op-amp 530 is a control value applied to thelimiting current source 404. In some implementations, the control valueis a function of the reference voltage and a voltage drop across theadjustable resistive load. In some implementations, the control value isa function of the difference between the reference voltage and thevoltage drop across the resistive load.

In some implementations, the adjustable load 540 includes a plurality ofparallel resistors 542 a, 542 b, 542 c, 542 d (i.e., R_(W1), R_(W2),R_(W3), R_(W4)) selectively connectable to a common node 544. Theadjustable load also includes a plurality of switches (or fuses) 543 a,543 b, 543 c, 543 d (i.e., F₁, F₂, F₃, F₄) coupled to selectivelyconnect one or more of the respective resistors to the common node 544in order to change the effective resistance of the adjustable resistiveload 540. To that end, the adjustable load 540 also includes a controlbus 541 coupled to the plurality of switches to provide respectivecontrol signals. In some implementations, each of the plurality ofresistors has a weighted value relative to at least one other of theplurality of resistors. In some implementations, the adjustableresistive load 540 is arranged in parallel with a fixed resistor 513(R₅). In operation, the resistors are selectively coupled to the commonnode 544 to trim the current provided by the limiting current source404.

FIG. 6 is a flowchart of an implementation of a method 600 of clampingthe current drawn by a power amplifier. In some implementations, method600 is performed by a power management system and/or controllerassociated with a power amplifier. Briefly, method 600 includes settingand applying a limiting electrical level to the control terminal of afirst transistor in order to set the operating mode of the firsttransistor. To that end, as represented by block 6-1, method 600includes producing a control value to compensate for temperature andmanufacturing process variations. For example, as represented by block6-1a and with further reference to FIG. 5, a control value can beproduce using an adjustable resistive load.

As represented by block 6-2, the method 600 includes generating alimiting electrical level using the control value. In someimplementations, as represented by block 6-2a, the method includesgenerating a limiting current level. As represented by block 6-3, themethod 600 includes applying the limiting electrical level to a controlterminal of a first transistor. In some implementations, as representedby block 6-3a, the method includes applying a limiting current level tothe base of a BJT.

FIG. 7 is a performance diagram showing a PA final stage beta versusload angle under current limits imposed by an implementation of the openloop current clamp circuit of FIG. 4. Specifically, FIG. 7 shows the PAfinal stage beta under current limit over temperature, supply, and loadangle. At high current load angles, beta maintains a high nominal valuewhich is desirable to reduce the area requirement for the clamp currentsource 404 of FIG. 4. The temperature characteristic of beta in thisoperation region is uniform and linear over temperature which reducesthe complexity of the temperature compensation circuitry.

FIG. 8 is a performance diagram showing a power amplifier final stagebeta versus temperature under current limits imposed by animplementation of the open loop current clamp circuit of FIG. 4. Ofparticular interest, FIG. 8 shows that has a negative temperature slope.In some implementations, to temperature compensate beta, I_(b) _(_)_(clamp) is linearly profiled with a positive temperature coefficientwith a magnitude equal to the temperature coefficient of beta. Duringnominal and low current conditions, final stage base current demand isless than I_(b) _(_) _(clamp). The current mirror forces the I_(b) _(_)_(clamp) clamp current source 404 into triode operation. The beta helpertransistor 410 (M₁) channel resistance increases causing the drainvoltage (V_(d)) to increase slightly below the respective supplyvoltage. Additionally, FIG. 9 shows the performance of the current clamp404 of FIG. 4 during the conditions described above.

FIGS. 10A-10C are schematic diagrams of different integrated circuitimplementations of the open loop current clamp circuit 420 of FIG. 4.While some example features are illustrated, those skilled in the artwill appreciate from the present disclosure that various other featureshave not been illustrated for the sake of brevity and so as not toobscure more pertinent aspects of the example implementations disclosedherein. To that end, for example, FIG. 10A shows that in someimplementations, some or all portions of the current clamp circuit 420can be part of a semiconductor die 1000. By way of an example, thecurrent clamp circuit 420 can be formed on a substrate 1002 of the die1000. A plurality of connection pads 1004 can also be formed on thesubstrate 1002 to facilitate functionalities associated with some or allportions of the current clamp circuit 420.

FIG. 10B shows that in some implementations, a semiconductor die 1000having a substrate 1002 can include some or all portions of the currentsource 404 and some or all portions of the current clamp circuit 420 ofFIG. 4. A plurality of connection pads 1004 can also be formed on thesubstrate 1002 to facilitate functionalities associated with some or allportions of the current source 404 and some or all portions of thecurrent clamp circuit 420 of FIG. 4.

FIG. 10C shows that in some embodiments, a semiconductor die 1000 havinga substrate 1002 can include some or all portions of the PA circuit 102,some or all portions of the current source 404 and some or all portionsof the current clamp circuit 420 of FIG. 4. A plurality of connectionpads 1004 can also be formed on the substrate 1002 to facilitatefunctionalities associated with some or all portions of the PA circuit102, the current source 404, and the current clamp circuit 420.

In some implementations, one or more features described herein can beincluded in a module. FIG. 11 is a schematic diagram of animplementation of a module 1100 including the open loop current clampcircuit 420 of FIG. 4. While some example features are illustrated,those skilled in the art will appreciate from the present disclosurethat various other features have not been illustrated for the sake ofbrevity and so as not to obscure more pertinent aspects of the exampleimplementations disclosed herein. The module 1100 includes a packagingsubstrate 1152, connection pads 1156, a CMOS (complementary metal oxidesemiconductor) die 1000, a HBT (heterojunction bipolar transistor) die1110, a matching network 108, and one or more surface mount devices1160.

The CMOS die 1000 includes a substrate 1002 including some or allportions of the current source 404 and some or all portions of thecurrent clamp circuit 420 of FIG. 4. A plurality of connection pads 1004is formed on the substrate 1002 to facilitate functionalities associatedwith some or all portions of the current source 404 and some or allportions of the current clamp circuit 420 of FIG. 4. Similarly, the HBTdie 1110 includes a substrate 1102 including some or all portions of thePA 102 and some or all portions of the bias circuitry provided to setthe quiescent conditions of the PA 102. The HBT die 1110 also includes aplurality of connection pads 1004 formed on the substrate 1102 tofacilitate functionalities associated with some or all portions of thePA 102 and some or all portions of the bias circuitry 1103.

The connection pads 1156 on the packaging substrate 1152 facilitateelectrical connections to and from each of the CMOS die 1000 and the HBTdie 1100. For example, the connection pads 1156 facilitate the use ofwirebonds 1154 for passing various signals and supply currents and/orvoltages to each of the CMOS die 1000 and the HBT die 1100.

In some implementations, the components mounted on the packagingsubstrate 1152 or formed on or in the packaging substrate 1152 canfurther include, for example, one or more surface mount devices (SMDs)(e.g., 1160) and one or more matching networks (e.g., 108). In someembodiments, the packaging substrate 1152 can include a laminatesubstrate.

In some implementations, the module 1100 can also include one or morepackaging structures to, for example, provide protection and facilitateeasier handling of the module 1100. Such a packaging structure caninclude an overmold formed over the packaging substrate 1152 anddimensioned to substantially encapsulate the various circuits andcomponents thereon.

It will be understood that although the module 1150 is described in thecontext of wirebond-based electrical connections, one or more featuresof the present disclosure can also be implemented in other packagingconfigurations, including flip-chip configurations.

In some implementations, a device and/or a circuit having one or morefeatures described herein can be included in an RF device such as awireless device. Such a device and/or a circuit can be implementeddirectly in the wireless device, in a modular form as described herein,or in some combination thereof. In some embodiments, such a wirelessdevice can include, for example, a cellular phone, a smart-phone, ahand-held wireless device with or without phone functionality, awireless tablet, a wireless router, a wireless access point, a wirelessbase station, etc. That is, those skilled in the art will alsoappreciate from the present disclosure that in various implementationsthe power amplifier open loop current clamp may be included in variousdevices, such as a computer, a laptop computer, a tablet device, anetbook, an internet kiosk, a personal digital assistant, an opticalmodem, a base station, a repeater, a wireless router, a mobile phone, asmartphone, a gaming device, a computer server, or any other computingdevice. In various implementations, such devices include one or moreprocessors, one or more types of memory, a display and/or other userinterface components such as a keyboard, a touch screen display, amouse, a track-pad, a digital camera and/or any number of supplementaldevices to add functionality.

FIG. 12 is a schematic diagram of an implementation of a wireless deviceincluding one or more features described herein, such as the open loopcurrent clamp circuit of FIG. 4. While some example features areillustrated, those skilled in the art will appreciate from the presentdisclosure that various other features have not been illustrated for thesake of brevity and so as not to obscure more pertinent aspects of theexample implementations disclosed herein.

One or more PAs 1216 as described herein are biased by respective biascircuit(s) (not shown) and compensated by respective compensationcircuit(s) (not shown). In some implementations the PAs 1216 arepackaged into a module including matching circuits 100. The PAs 1216 canreceive respective RF signals from a transceiver 1214, that can beconfigured and operated in known manners to generate RF signals to beamplified and transmitted, and to process received signals. Thetransceiver 1214 is shown to interact with a baseband sub-system 1210that is configured to provide conversion between data and/or voicesignals suitable for a user and RF signals suitable for the transceiver1214. The transceiver 1214 is also shown to be connected to a powermanagement component 1206 that is configured to manage power for theoperation of the wireless device 1200. Such power management can alsocontrol operations of the baseband sub-system 1210, the current clamp1208 coupled to the PAs 1216, and access to the battery 1209.

The baseband sub-system 1210 is shown to be connected to a userinterface 1202 to facilitate various input and output of voice and/ordata provided to and received from the user. The baseband sub-system1210 can also be connected to a memory 1204 that is configured to storedata and/or instructions to facilitate the operation of the wirelessdevice, and/or to provide storage of information for the user.

In the example wireless device 1200, outputs of the PAs 1216 are shownto be matched and routed to an antenna 1224 via respective duplexers1220 and a band-selection switch 1222. The band-selection switch 1222can include, for example, a single-pole-multiple-throw (e.g., SP4T)switch to allow selection of an operating band (e.g., Band 2). In someembodiments, each duplexer 1220 can allow transmit and receiveoperations to be performed simultaneously using a common antenna (e.g.,1224). In FIG. 17, received signals are shown to be routed to “Rx” paths(not shown) that can include, for example, a low-noise amplifier (LNA).

A number of other wireless device configurations can utilize one or morefeatures described herein. For example, a wireless device does not needto be a multi-band device. In another example, a wireless device caninclude additional antennas such as diversity antenna, and additionalconnectivity features such as Wi-Fi, Bluetooth, and GPS.

While various aspects of implementations within the scope of theappended claims are described above, it should be apparent that thevarious features of implementations described above may be embodied in awide variety of forms and that any specific structure and/or functiondescribed above is merely illustrative. Based on the present disclosureone skilled in the art should appreciate that an aspect described hereinmay be implemented independently of any other aspects and that two ormore of these aspects may be combined in various ways. For example, anapparatus may be implemented and/or a method may be practiced using anynumber of the aspects set forth herein. In addition, such an apparatusmay be implemented and/or such a method may be practiced using otherstructure and/or functionality in addition to or other than one or moreof the aspects set forth herein.

It will also be understood that, although the terms “first,” “second,”etc. may be used herein to describe various elements, these elementsshould not be limited by these terms. These terms are only used todistinguish one element from another. For example, a first contact couldbe termed a second contact, and, similarly, a second contact could betermed a first contact, which changing the meaning of the description,so long as all occurrences of the “first contact” are renamedconsistently and all occurrences of the second contact are renamedconsistently. The first contact and the second contact are bothcontacts, but they are not the same contact.

The terminology used herein is for the purpose of describing particularembodiments only and is not intended to be limiting of the claims. Asused in the description of the embodiments and the appended claims, thesingular forms “a”, “an” and “the” are intended to include the pluralforms as well, unless the context clearly indicates otherwise. It willalso be understood that the term “and/or” as used herein refers to andencompasses any and all possible combinations of one or more of theassociated listed items. It will be further understood that the terms“comprises” and/or “comprising,” when used in this specification,specify the presence of stated features, integers, steps, operations,elements, and/or components, but do not preclude the presence oraddition of one or more other features, integers, steps, operations,elements, components, and/or groups thereof.

As used herein, the term “if” may be construed to mean “when” or “upon”or “in response to determining” or “in accordance with a determination”or “in response to detecting,” that a stated condition precedent istrue, depending on the context. Similarly, the phrase “if it isdetermined [that a stated condition precedent is true]” or “if [a statedcondition precedent is true]” or “when [a stated condition precedent istrue]” may be construed to mean “upon determining” or “in response todetermining” or “in accordance with a determination” or “upon detecting”or “in response to detecting” that the stated condition precedent istrue, depending on the context.

What is claimed is:
 1. An open loop current clamp to restrict thecurrent draw of a first transistor, the open loop current clampcomprising: a current trim module including a voltage source having avoltage output terminal to provide a reference voltage, an adjustableresistive load and a control value generator configured to provide acontrol value as a function of the reference voltage and a voltage dropacross the adjustable resistive load; and a limiting first currentsource having respective input and output terminals, the input terminalcoupled to the current trim module to receive the control value, and theoutput terminal coupled to a first control terminal of the firsttransistor to provide a limiting electrical level produced in responseto the control value, such that the limiting electrical levelsubstantially sets a first mode of operation for the first transistorsuch that the current draw of the first transistor is substantiallydetermined by the first mode of operation and the limiting electricallevel such that a voltage at a first output terminal of the firsttransistor exerts reduced influence on the current draw.
 2. The openloop current clamp of claim 1 further comprising a current mirrorincluding a second transistor, the second transistor having a secondcontrol terminal and a second input terminal, the second controlterminal of the second transistor coupled to the first control terminalof the first transistor, and the current mirror also including a secondcurrent source coupled to the second input terminal of the secondtransistor to form an input portion of the current mirror, the secondcurrent source provided to set the nominal DC current drawn by the inputportion of the current mirror, and an output portion of the currentmirror includes the first transistor.
 3. The open loop current clamp ofclaim 2 further comprising a resistor and capacitor in series betweenthe second input terminal and second control terminal of the secondtransistor.
 4. The open loop current clamp of claim 2 wherein the firstand second transistors are together characterized by a device area ratiothat is used to determine in part the current drawn by the firsttransistor relative to the output of the second current source coupledto the second transistor.
 5. The open loop current clamp of claim 2further comprising a gain helper having first, second and thirdterminals, wherein the first terminal is coupled to the output of thesecond current source, the second terminal is coupled to the output ofthe limiting first current source, and the third terminal is coupled tothe first control terminal of the first transistor.
 6. The open loopcurrent clamp of claim 5 wherein the gain helper is a transistor.
 7. Theopen loop current clamp of claim 6 wherein the first and secondtransistors are bipolar junction transistors, the gain helper is a metaloxide semiconductor transistor, and the gain is defined as the currentgain of a bipolar junction transistor.
 8. The open loop current clamp ofclaim 1 wherein the first transistor comprises a bipolar junctiontransistor, and the limiting electrical level is a current provided tothe base of the bipolar junction transistor.
 9. The open loop currentclamp of claim 1 wherein the control value is a function of thedifference between the reference voltage and the voltage drop across theresistive load.
 10. The open loop current clamp of claim 1 wherein theadjustable resistive load comprises: a plurality of parallel resistorsselectively connectable to a common node; a plurality of switchescoupled to selectively connect one or more of the respective resistorsto the common node in order to change the effective resistance of theadjustable resistive load; and a control bus coupled to the plurality ofswitches to provide respective control signals.
 11. The open loopcurrent clamp of claim 1 wherein the first transistor is a bipolarjunction transistor.
 12. The open loop current clamp of claim 1 whereinthe first transistor is heterojunction bipolar transistor.
 13. The openloop current clamp of claim 1 wherein the limiting first current sourceand current trim module are included on a first die that is separatefrom a second die including the first transistor.
 14. The open loopcurrent clamp of claim 1 wherein the limiting first current source andcurrent trim module are included on the same die as the firsttransistor.
 15. The open loop current clamp of claim 1 wherein thelimiting first current source includes an adjustable current mirror, theadjustable current mirror comprising: a plurality of paralleltransistors selectively connectable to a common node; a plurality ofswitches coupled to selectively connect one or more of the respectivetransistors to the common node in order set a reference current of theadjustable current mirror; and a control bus coupled to the plurality ofswitches to provide respective control signals.
 16. A power amplifiermodule comprising: a packaging substrate configured to receive aplurality of components; a power amplifier circuit provided in a dieincluded on the packaging substrate, the power amplifier including afirst transistor having a first control terminal and a first outputterminal; a current trim module including a voltage source having avoltage output terminal to provide a reference voltage, an adjustableresistive load and a control value generator configured to provide acontrol value as a function of the reference voltage and a voltage dropacross the adjustable resistive load; and a limiting first currentsource having respective input and output terminals, the input terminalcoupled to the current trim module to receive the control value, and theoutput terminal coupled to the first control terminal of the firsttransistor to provide a limiting electrical level produced in responseto the control value, such that the limiting electrical levelsubstantially sets a first mode of operation for the first transistorsuch that the current draw of the first transistor is substantiallydetermined by the first mode of operation and the limiting electricallevel such that a voltage at the first output terminal of the firsttransistor exerts reduced influence on the current draw.
 17. The poweramplifier module of claim 16 wherein at least a portion of one of thecurrent trim module and limiting first current source is provided on thesame die as the power amplifier.
 18. The power amplifier module of claim17 wherein the die is a heterojunction bipolar junction transistor die.19. A radio frequency (RF) device comprising: a transceiver configuredto process RF signals; an antenna in communication with the transceiverconfigured to facilitate transmission of an amplified RF signal; and apower amplifier module connected to the transceiver and configured togenerate the amplified RF signal, the power amplifier module including afirst transistor, a current trim module including a voltage sourcehaving a voltage output terminal to provide a reference voltage, anadjustable resistive load and a control value generator configured toprovide a control value as a function of the reference voltage and avoltage drop across the adjustable resistive load, and a limiting firstcurrent source having respective input and output terminals, the inputterminal coupled to the current trim module to receive the controlvalue, and the output terminal coupled to a first control terminal ofthe first transistor to provide a limiting electrical level produced inresponse to the control value, such that the limiting electrical levelsubstantially sets a first mode of operation for the first transistorsuch that the current draw of the first transistor is substantiallydetermined by the first mode of operation and the limiting electricallevel such that a voltage at a first output terminal of the firsttransistor exerts reduced influence on the current draw.
 20. The radiofrequency (RF) device of claim 19 wherein the adjustable resistive loadcomprises: a plurality of parallel resistors selectively connectable toa common node; a plurality of switches coupled to selectively connectone or more of the respective resistors to the common node in order tochange the effective resistance of the adjustable resistive load; and acontrol bus coupled to the plurality of switches to provide respectivecontrol signals.